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MIC2171
Micrel
MIC2171
100kHz 2.5A Switching Regulator Preliminary Information
General Description
The MIC2171 is a complete 100kHz SMPS current-mode controller with an internal 65V 2.5A power switch. Although primarily intended for voltage step-up applications, the floating switch architecture of the MIC2171 makes it practical for step-down, inverting, and Cuk configurations as well as isolated topologies. Operating from 3V to 40V, the MIC2171 draws only 7mA of quiescent current, making it attractive for battery operated supplies. The MIC2171 is available in a 5-pin TO-220 or TO-263 for -40C to +85C operation.
Features
* * * * * * * * * * * * * * 2.5A, 65V internal switch rating 3V to 40V input voltage range Current-mode operation, 2.5A peak Internal cycle-by-cycle current limit Thermal shutdown Twice the frequency of the LM2577 Low external parts count Operates in most switching topologies 7mA quiescent current (operating) Fits LT1171/LM2577 TO-220 and TO-263 sockets
Applications
Laptop/palmtop computers Battery operated equipment Hand-held instruments Off-line converter up to 50W (requires external power switch) * Predriver for higher power capability
4
Typical Applications
+5V (4.75V min.) L1 15H IN SW MIC2171 COMP R3 1k GND C3 1F FB 1N5822 D1
C1* 47F VOUT +12V, 0.25A R1 10.7k 1% C2 470F R2 1.24k 1%
VIN 4V to 6V
T1 R4* C3* D2 1N5818 C4 470F
VOUT 5V, 0.5A R1 3.74k 1%
C1 47F
IN SW MIC2171 COMP R3 1k C2 1F GND FB
D1* 1:1.25 LPRI = 12H
R2 1.24k 1%
* Locate near MIC2171 when supply leads > 2"
* Optional voltage clipper (may be req'd if T1 leakage inductance too high)
Figure 1. MIC2171 5V to 12V Boost Converter
Figure 2. MIC2171 5V Flyback Converter
1997
4-3
MIC2171
Micrel
Ordering Information
Part Number MIC2171BT MIC2171BU Temperature Range -40C to +85C -40C to +85C Package 5-lead TO-220 5-lead TO-263
Pin Configuration
5 4 3 2 1 Tab GND
IN SW GND FB COMP Tab GND
5 4 3 2 1
IN SW GND FB COMP
5-lead TO-220 (BT)
5-lead TO-263 (BU)
Pin Description
Pin Number 1 Pin Name COMP Pin Function Frequency Compensation: Output of transconductance-type error amplifier. Primary function is for loop stabilization. Can also be used for output voltage soft-start and current limit tailoring. Feedback: Inverting input of error amplifier. Connect to external resistive divider to set power supply output voltage. Ground: Connect directly to the input filter capacitor for proper operation (see applications info). Power Switch Collector: Collector of NPN switch. Connect to external inductor or input voltage depending on circuit topology. Supply Voltage: 3.0V to 40V
2 3 4 5
FB GND SW IN
4-4
1997
MIC2171
Micrel
Junction Temperature ................................ -55C to 150C Thermal Resistance JA 5-lead TO-220, Note 1................................. 45C/W JA 5-lead TO-263, Note 2................................. 45C/W Storage Temperature ............................... -65C to +150C Soldering (10 sec.) .................................................. +300C
Absolute Maximum Ratings
Input Voltage (VIN) ........................................................ 40V Switch Voltage (VSW) .................................................... 65V Feedback Voltage (transient, 1ms) (VFB) ................... 15V Operating Temperature Range ...................... -40 to +85C
Electrical Characteristics
VIN = 5V; TA = 25C, bold values indicate -40C TA +85C; unless noted. Parameter Reference Section Feedback Voltage (VFB) Feedback Voltage Line Regulation Feedback Bias Current (IFB) Error Amplifier Section Transconductance (gm) Voltage Gain (AV) Output Current Output Swing Compensation Pin Threshold Output Switch Section ON Resistance Current Limit ISW = 2A, VFB = 0.8V Duty Cycle = 50%, TJ 25C Duty Cycle = 50%, TJ < 25C Duty Cycle = 80%, Note 3 3V VIN 40V ISW = 5mA 2.5 2.5 2.0 65 0.37 3.6 4.0 3.0 75 0.50 0.55 5 5.5 5 A A A V ICOMP = 25A 0.9V VCOMP 1.4V VCOMP = 1.5V High Clamp, VFB = 1V Low Clamp, VFB = 1.5V Duty Cycle = 0 3.0 2.4 400 125 100 1.8 0.25 0.8 0.6 3.9 800 175 2.1 0.35 0.9 6.0 7.0 2000 350 400 2.3 0.52 1.08 1.25 A/mV A/mV V/V A A V V V V VCOMP = 1.24V 3V VIN 40V VCOMP = 1.24V VFB = 1.24V 1.220 1.214 1.240 .06 310 750 1100 1.264 1.274 V V %/V nA nA Conditions Min Typ Max Units
4
Breakdown Voltage (BV) Oscillator Section Frequency (fO) Duty Cycle [(max)] Input Supply Voltage Section Minimum Operating Voltage Quiescent Current (IQ) Supply Current Increase (IIN)
Note 1 Note 2 Note 3
88 85 80
100 90
112 115 95
kHz kHz %
2.7 3V VIN 40V, VCOMP = 0.6V, ISW = 0 ISW = 2A, VCOMP = 1.5V, during swich on-time 7 9
3.0 9 20
V mA mA
General Note Devices are ESD sensitive. Handling precautions required. Mounted vertically, no external heat sink, 1/4 inch leads soldered to PC board containing approximently 4 inch squared copper area surrounding leads. All ground leads soldered to approximently 2 inches squared of horizontal PC board copper area. For duty cycles () between 50% and 95%, minimum guaranteed switch current is ICL = 1.66 (2-) Amp (Pk).
1997
4-5
MIC2171
Micrel
Typical Performance Characteristics
Minimum Operating Voltage
Minimum Operating Voltage (V) 2.9
Feedback Bias Current
Feedback Voltage Change (mV)
Feedback Voltage Line Regulation
5 4 3 2 1 0 -1 -2 -3 -4 -5 0 TJ = -40C 10 20 30 VIN Operating (V) 40 TJ = 125C
800 Feedback Bias Current (nA) 700 600 500 400 300 200 100 0 -100 -50 0 50 100 Temperature (C) 150
2.8 2.7 2.6 Switch Current = 2A 2.5 2.4 2.3 -100 -50 0 50 100 Temperature (C) 150
TJ = 25C
Supply Current
Average Supply Current (mA)
15 14 Supply Current (mA) 13 12 11 10 9 8 7 6 5 D.C. = 90%
Supply Current
50 10 Supply Current (mA) 9 8 7 6 5 4 3 2 1 0 -100
Supply Current
VCOMP = 0.6V
ISW = 0
40 30 20 = 50% 10 0 = 90%
D.C. = 50% D.C. = 0%
0
10 20 30 VIN Operating Voltage (V)
40
0
1 2 3 Switch Current (A)
4
-50 0 50 100 Temperature (C)
150
1.6 1.4 Switch ON Voltage (V)
Switch On-Voltage
Oscillator Frequency
120 110 100 90 80 70 3 60 -50 0 50 100 Temperature (C) 150
Current Limit
8
1.0 0.8 0.6 0.4 0.2 0 0
TJ = -40C
Switch Current (A)
Frequency (kHz)
1.2
TJ = 25C
6 -40C 25C
4
TJ = 125C
2
125C
1 2 Switch Current (A)
0
0
20
40 60 80 Duty Cycle (%)
100
Error Amplifier Gain
Transconductance (A/mV) 5.0 4.5 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0.0 -100 -50 0 50 100 Temperature (C) 150 Transconductance (S) 7000 6000 5000
Error Amplifier Gain
-30 0 30 Phase Shift () 60 90 120 150 1000 0 1 10 100 1000 Frequency (kHz) 10000 180 210 1
Error Amplifier Phase
4000 3000 2000
10
100 1000 Frequency (kHz)
10000
4-6
1997
MIC2171
Micrel
Block Diagram MIC2171
IN
Reg.
2.3V Anti-Sat. 100kHz Osc. Logic Driver
D1 SW
Q1
FB Error Amp.
Comparator Current Amp.
1.24V Ref.
COMP
GND
4
Functional Description
Refer to "Block Diagram MIC2171". Internal Power The MIC2171 operates when VIN is 2.6V. An internal 2.3V regulator supplies biasing to all internal circuitry including a precision 1.24V band gap reference. PWM Operation The 100kHz oscillator generates a signal with a duty cycle of approximately 90%. The current-mode comparator output is used to reduce the duty cycle when the current amplifier output voltage exceeds the error amplifier output voltage. The resulting PWM signal controls a driver which supplies base current to output transistor Q1. Current-Mode Advantages The MIC2171 operates in current mode rather than voltage mode. There are three distinct advantages to this technique. Feedback loop compensation is greatly simplified because inductor current sensing removes a pole from the closed loop
response. Inherent cycle-by-cycle current limiting greatly improves the power switch reliability and provides automatic output current limiting. Finally, current-mode operation provides automatic input voltage feed forward which prevents instantaneous input voltage changes from disturbing the output voltage setting. Anti-Saturation The anti-saturation diode (D1) increases the usable duty cycle range of the MIC2171 by eliminating the base to collector stored charge which would delay Q1's turnoff. Compensation Loop stability compensation of the MIC2171 can be accomplished by connecting an appropriate network from either COMP to circuit ground (see typical Applications) or COMP to FB. The error amplifier output (COMP) is also useful for soft start and current limiting. Because the error amplifier output is a transconductance type, the output impedance is relatively high which means the output voltage can be easily clamped or adjusted externally.
1997
4-7
MIC2171
Micrel
The device operating losses are the dc losses associated with biasing all of the internal functions plus the losses of the power switch driver circuitry. The dc losses are calculated from the supply voltage (VIN) and device supply current (IQ). The MIC2171 supply current is almost constant regardless of the supply voltage (see "Electrical Characteristics"). The driver section losses (not including the switch) are a function of supply voltage, power switch current, and duty cycle.
P(bias+driver) = VIN IQ + VIN(min) x ISW x IIN
MIC2171 COMP D1 D2 R1 C1 C2
Applications Information
Soft Start A diode-coupled capacitor from COMP to circuit ground slows the output voltage rise at turn on (Figure 3).
VIN IN
(
)(
)
where: P(bias+driver) = device operating losses VIN(min) = supply voltage = VIN - VSW IQ = typical quiescent supply current ICL = power switch current limit IIN = typical supply current increase As a practical example refer to Figure 1. VIN = 5.0V IQ = 0.007A ICL = 2.21A = 66.2% (0.662) Then: VIN(min) = 5 - (2.21 x 0.37) = 4.18V
Figure 3. Soft Start The additional time it takes for the error amplifier to charge the capacitor corresponds to the time it takes the output to reach regulation. Diode D1 discharges C1 when VIN is removed. Current Limit
VIN
IN SW MIC2171 FB GND COMP VOUT
P(bias
+ driver)
= (5 x 0.007) + (4.18 x 2.21 x .009)
R1 Q1 C1 R2
R3 C2
ICL 0.6V/R2 Note: Input and output returns not common.
P(bias+driver) = 0.1W Power switch dissipation calculations are greatly simplified by making two assumptions which are usually fairly accurate. First, the majority of losses in the power switch are due to on-losses. To find these losses, assign a resistance value to the collector/emitter terminals of the device using the saturation voltage versus collector current curves (see Typical Performance Characteristics). Power switch losses are calculated by modeling the switch as a resistor with the switch duty cycle modifying the average power dissipation. PSW = (ISW)2 RSW where: = duty cycle
= VOUT + VF - VIN(min) VOUT + VF
Figure 4. Current Limit The maximum current limit of the MIC2171 can be reduced by adding a voltage clamp to the COMP output (Figure 4). This feature can be useful in applications requiring either a complete shutdown of Q1's switching action or a form of current fold-back limiting. This use of the COMP output does not disable the oscillator, amplifiers or other circuitry, therefore the supply current is never less than approximately 5mA. Thermal Management Although the MIC2171 family contains thermal protection circuitry, for best reliability, avoid prolonged operation with junction temperatures near the rated maximum. The junction temperature is determined by first calculating the power dissipation of the device. For the MIC2171, the total power dissipation is the sum of the device operating losses and power switch losses.
VSW = ICL (RSW) VOUT = output voltage VF = D1 forward voltage drop at IOUT From the Typical performance Characteristics: RSW = 0.37 Then: PSW = (2.21)2 x 0.37 x 0.662 PSW) = 1.2W P(total) = 1.2 + 0.1 P(total) = 1.3W 4-8 1997
MIC2171
The junction temperature for any semiconductor is calculated using the following: TJ = TA + P(total) JA Where: TJ = junction temperature TA = ambient temperature (maximum) P(total) = total power dissipation JA = junction to ambient thermal resistance For the practical example: TA = 70C JA = 45C/W (TO-220) Then: TJ = 70 + (1.24 x 45) TJ = 126C This junction temperature is below the rated maximum of 150C. Grounding Refer to Figure 5. Heavy lines indicate high current paths.
VIN IN SW MIC2171 FB COMP
Micrel
mode is preferred because the feedback control of the converter is simpler. When L1 discharges its current completely during the MIC2171 off-time, it is operating in discontinuous mode. L1 is operating in continuous mode if it does not discharge completely before the MIC2171 power switch is turned on again.
Discontinuous Mode Design
Given the maximum output current, solve equation (1) to determine whether the device can operate in discontinuous mode without initiating the internal device current limit. ICL V 2 IN(min) VOUT
(1)
IOUT
(1a)
=
VOUT + VF - VIN(min) VOUT + VF
Where: ICL = internal switch current limit ICL = 2.5A when < 50% ICL = 1.67 (2 - ) when 50% (Refer to Electrical Characteristics.) IOUT = maximum output current VIN(min) = minimum input voltage = VIN - VSW = duty cycle VOUT = required output voltage VF = D1 forward voltage drop For the example in Figure 1.
4
GND
Single point ground
Figure 5. Single Point Ground A single point ground is strongly recommended for proper operation. The signal ground, compensation network ground, and feedback network connections are sensitive to minor voltage variations. The input and output capacitor grounds and power ground conductors will exhibit voltage drop when carrying large currents. Keep the sensitive circuit ground traces separate from the power ground traces. Small voltage variations applied to the sensitive circuits can prevent the MIC2171 or any switching regulator from functioning properly. Boost Conversion Refer to Figure 1 for a typical boost conversion application where a +5V logic supply is available but +12V at 0.25A is required. The first step in designing a boost converter is determining whether inductor L1 will cause the converter to operate in either continuous or discontinuous mode. Discontinuous 1997 4-9 Then:
IOUT = 0.25A ICL = 1.67 (2-0.662) = 2.24A VIN(min) = 4.18V = 0.662 VOUT = 12.0V VF = 0.36V (@ .26A, 70C)
2.235 x 4.178 x 0.662 2 12
IOUT
IOUT 0.258A This value is greater than the 0.25A output current requirement, so we can proceed to find the minimum inductance value of L1 for discontinuous operation at POUT. (2)
L1
(VIN )2
2 POUT fSW
Where: POUT = 12 x 0.25 = 3W fSW = 1x105Hz (100kHz)
MIC2171
For our practical example:
Micrel
down (failure) of the MIC2171's internal power switch.
2
x 0.662) 2 x 3.0 x 1x 105 L1 12.4H (use 15H) Equation (3) solves for L1's maximum current value. L1
(3)
IL1(peak) = VIN T ON L1
(4.178
Discontinuous Mode Design
When designing a discontinuous flyback converter, first determine whether the device can safely handle the peak primary current demand placed on it by the output power. Equation (8) finds the maximum duty cycle required for a given input voltage and output power. If the duty cycle is greater than 0.8, discontinuous operation cannot be used. (8)
Where: TON = / fSW = 6.62x10-6 sec
4.178 x 6.62 x 10-6 15 x 10-6 IL1(peak) = 1.84A IL1(peak) =
ICL VIN(min) - VSW
(
2 POUT
)
For a practical example let: (see Figure 2) POUT = 5.0V x 0.5A = 2.5W VIN = 4.0V to 6.0V ICL = 2.5A when < 50% 1.67 (2 - ) when 50% Then:
VIN(min) = VIN - ICL x RSW
Use a 15H inductor with a peak current rating of at least 2A. Flyback Conversion Flyback converter topology may be used in low power applications where voltage isolation is required or whenever the input voltage can be less than or greater than the output voltage. As with the step-up converter the inductor (transformer primary) current can be continuous or discontinuous. Discontinuous operation is recommended. Figure 2 shows a practical flyback converter design using the MIC2171.
(
)
VIN(min) = 4 - 0.78V VIN(min) = 3.22V 0.74 (74%), less than 0.8 so discontinous is permitted. A few iterations of equation (8) may be required if the duty cycle is found to be greater than 50%. Calculate the maximum transformer turns ratio a, or NPRI/NSEC, that will guarantee safe operation of the MIC2171 power switch. (9)
a V CE FCE - VIN(max) V SEC
Switch Operation
During Q1's on time (Q1 is the internal NPN transistor--see block diagrams), energy is stored in T1's primary inductance. During Q1's off time, stored energy is partially discharged into C4 (output filter capacitor). Careful selection of a low ESR capacitor for C4 may provide satisfactory output ripple voltage making additional filter stages unnecessary. C1 (input capacitor) may be reduced or eliminated if the MIC2171 is located near a low impedance voltage source.
Output Diode
The output diode allows T1 to store energy in its primary inductance (D2 nonconducting) and release energy into C4 (D2 conducting). The low forward voltage drop of a Schottky diode minimizes power loss in D2.
Frequency Compensation
A simple frequency compensation network consisting of R3 and C2 prevents output oscillations. High impedance output stages (transconductance type) in the MIC2171 often permit simplified loop-stability solutions to be connected to circuit ground, although a more conventional technique of connecting the components from the error amplifier output to its inverting input is also possible.
Where: a = transformer maximum turns ratio VCE = power switch collector to emitter maximum voltage FCE = safety derating factor (0.8 for most commercial and industrial applications) VIN(max) = maximum input voltage VSEC = transformer secondary voltage (VOUT + VF) For the practical example: VCE = 65V max. for the MIC2171 FCE = 0.8 VSEC = 5.6V Then:
a 65 x 0.8 - 6.0 5.6
Voltage Clipper
Care must be taken to minimize T1's leakage inductance, otherwise it may be necessary to incorporate the voltage clipper consisting of D1, R4, and C3 to avoid second break-
a 8.2 (NPRI/NSEC) Next, calculate the maximum primary inductance required to store the needed output energy with a power switch duty cycle of 55%. 1997
4-10
MIC2171
(10) LPRI 0.5 fSW VIN(min)2 TON2 POUT (12) Then: a LPRI L SEC
Micrel
Where: LPRI = maximum primary inductance fSW = device switching frequency (100kHz) VIN(min) = minimum input voltage TON = power switch on time Then:
LPRI 0.5 x 1x 105 x (3.22) 2.5
2
a
11.4 = 1.20 7.9
x 7.4 x 10-6
(
)2
This ratio is less than the ratio calculated in equation (9). When specifying the transformer it is necessary to know the primary peak current which must be withstood without saturating the transformer core. (13) So:
IPEAK(pri) = VIN(min) T ON LPRI
LPRI 11.4H Use an 12H primary inductance to overcome circuit inefficiencies. To complete the design the inductance value of the secondary is found which will guarantee that the energy stored in the transformer during the power switch on time will be completed discharged into the output during the off-time. This is necessary when operating in discontinuous-mode. (11) L SEC 0.5 f SW V SEC 2 T OFF 2 POUT
IPEAK(pri) =
3.22 x 7.6 x 10-6 12H
IPEAK(pri) = 2.1A Now find the minimum reverse voltage requirement for the output rectifier. This rectifier must have an average current rating greater than the maximum output current of 0.5A. VBR VIN(max) + V OUT a FBR a
(14)
(
)
4
Where: LSEC = maximum secondary inductance TOFF = power switch off time Then:
L SEC 0.5 x 1x 105 x (5.41) 2.5
2
Where: VBR = output rectifier maximum peak reverse voltage rating a = transformer turns ratio (1.2) FBR = reverse voltage safety derating factor (0.8) Then:
VBR 6.0 + (5.0 x 1.2) 0.8 x 1.2
x 2.6 x 10-6
(
)2
LSEC 7.9H Finally, recalculate the transformer turns ratio to insure that it is less than the value earlier found in equation (9).
VBR 12.5V A 1N5817 will safely handle voltage and current requirements in this example.
1997
4-11
MIC2171
Forward Converters Micrel's MIC2171 can be used in several circuit configurations to generate an output voltage which is less than the input voltage (buck or step-down topology). Figure 7 shows the MIC2171 in a voltage step-down application. Because of the internal architecture of these devices, more external components are required to implement a step-down regulator than with other devices offered by Micrel (refer to the LM257x or MIC457x family of buck switchers). However, for step-down conversion requiring a transformer (forward), the MIC2171 is a good choice. A 12V to 5V step-down converter using transformer isolation (forward) is shown in Figure 7. Unlike the isolated flyback converter which stores energy in the primary inductance during the controller's on-time and releases it to the load during the off-time, the forward converter transfers energy to the output during the on-time, using the off-time to reset the transformer core. In the application shown, the transformer
Micrel
core is reset by the tertiary winding discharging T1's peak magnetizing current through D2. For most forward converters the duty cycle is limited to 50%, allowing the transformer flux to reset with only two times the input voltage appearing across the power switch. Although during normal operation this circuit's duty cycle is well below 50%, the MIC2172 has a maximum duty cycle capability of 90%. If 90% was required during operation (start-up and high load currents), a complete reset of the transformer during the off-time would require the voltage across the power switch to be ten times the input voltage. This would limit the input voltage to 6V or less for forward converter applications. To prevent core saturation, the application given here uses a duty cycle limiter consisting of Q1, C4 and R3. Whenever the MIC2171 exceeds a duty cycle of 50%, T1's reset winding current turns Q1 on. This action reduces the duty cycle of the MIC2171 until T1 is able to reset during each cycle.
T1 1:1:1 VIN 12V R1* C2* D1* SW MIC2171 C1 22F GND FB COMP R2 1k C3 1F D2 1N5819 Q1 R3 C4
D3 1N5819
L1 100H
D4 1N5819
R4 C5 3.74k 470F 1%
VOUT 5V, 1A
IN
R5 1.24k 1%
* Voltage clipper Duty cycle limiter
Figure 7. MIC2171 Forward Converter
4-12
1997

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